Wireless data communication method via ultra-wide band encoded data signals, and receiver device for implementing the same

ABSTRACT

This invention concerns a wireless data communication method, wherein a transmitter device having a first wide band antenna transmits ultra-wide band encoded data signals to a receiver device having a second wide band antenna for receiving the direct and/or multiple path coded data signals. The transmitted data is defined by one or several sequences of N pulses where N is an integer number greater than 1. The arrangement of the N pulses of each sequence represents a data coding relative to the transmitter device. The N pulses of a sequence of direct and/or multiple path coded data signals received by the receiver device are processed each in one among N corresponding reception time windows. Each of the N reception time windows is positioned in time based on a known theoretic arrangement of the N pulses of signals transmitted by the transmitter device. An adding operation of the N windows is then performed in the receiver device so that the amplitude level of the constantly added pulses is higher than the amplitude level of the noise sensed by the receiver device.

The present invention concerns a method for the wireless communicationof data between a transmitter device and a receiver device. Thetransmitter device transmits ultra-wide band encoded data signals via afirst wide band antenna, and the receiver device receives direct and/ormultiple path encoded data signals via a second wide band antenna fromthe transmitter device. The transmitted data is defined by one orseveral successive sequences of N pulses where N is an integer numbergreater than 1. The arrangement of the N pulses of each sequencerepresents data encoding relating to the transmitter device, i.e.personalising the transmitter device.

The invention also concerns the receiver device for implementing themethod.

In the present description, “data” means textual information, whichincludes one or several symbols or characters, audiovisual information,synchronisation information or positioning information or otherinformation. The data transmitted in the data signals is defined by oneor several very short pulse sequences whose encoding can be defined bythe time difference between each pulse.

Ultra-wide band data transmission technology is achieved using datasignals which include a series of very short pulses without using acarrier frequency. The width of these pulses can be less than 1 ns.Since the data signal pulses are very short in the time domain, whenconverting into the frequency domain, this leads to an ultra-wide bandspectrum, which defines UWB technology. The frequency spectrum can rangefrom 500 MHz to several GHz. The frequency bandwidth is generallygreater than 25% in relation to the central frequency for ultra-wideband technology.

Data transmission via ultra-wide band technology normally occurs from ashort distance with low transmitted pulse power. This is generally dueto the fact that the frequency spectrum is shared with narrow bandtransmission devices. This means that a single pulse is generallyreceived with a lower power level than the noise level. Thus, it isoften necessary to use more than one energy pulse to transmit a singlesymbol or character in order for it to be recognised by the receiverdevice.

For the transmission of encoded data signals which includes one or moresuccessive sequences of N very short pulses, the pulses can be ofdifferent shapes provided that their width is generally less than 1 ns.They may be, for example, Gaussian shape pulses with one or twopolarities or alternations.

Since several ultra-wide band (UWB) transmitter and receiver devices canbe located in proximity in the same space for transmitting data signals,as a rule the transmitted data signal sequence encoding is personalisedfor the transmitter device. In this way, the receiver device canrecognise the encoded signals from a particular transmitter device. Inaddition, all of the codes used for encoding data are, as a rule,orthogonal, which means that when they are correlated with each other,the correlation result gives a value close to 0.

Usually, the data transmitted in pulse sequence signals can be encodedfor example by pulse position modulation (PPM). The time differencebetween each pulse, and the instant that the first pulse of eachsequence appears can thus define the desired encoding for datacommunication. In order to do this, the pulses of each sequence aretransmitted at a pulse repetition frequency (PRF), which can be greaterfor example than 10 MHz. Each of the pulses is thus transmitted in arepetition window of a determined length which can be for example 20 nsor more. As a function of the desired time encoding, the pulse may belead or lag in relation to a determined theoretical transmissionposition so as to be able to code for example a “0” or a “1”.

When a pulse sequence encoded signal transmission is carried out asabove-mentioned, it is necessary that the pulses can be detected as afunction of their position via a PPM modulation for the signal receptionin the receiver device. This generally requires a high time coherence intransmitter and receiver devices for the detection of transmitted data.

The encoded data signals, which are transmitted by the transmitterdevice, can be reflected or refracted by various obstacles before beingpicked up by the receiver device. Consequently, several time shiftedencoded signals, i.e. direct and/or multiple path signals, which includeidentical data, can be picked up by the receiver device.

Several techniques for demodulating the information contained in encodeddata signals received by a conventional receiver device have alreadybeen proposed in the past. One of these known techniques consists incorrelating encoded data signals picked up and shaped in the receiverdevice with a reference signal lead replica and lag replica. Thecorrelated phase lead and phase lag signals are then integrated, and acode adjustment is made of each replica in a code control loop until thelevel of the integrated phase lead and phase lag signals is identical.However, if all of the multiple path signals have to be detected,several correlation stages are used in parallel. Consequently, theelectric power consumption of the receiver device is large, and manyelectronic components are necessary for processing signals in thereceiver device, which constitutes a major drawback.

US Patent Application No. 2003/0095609 discloses a UWB method andapparatus for receiving several time spaced signals. The ultra-wide bandsignals are received by an antenna of the apparatus in order to becorrelated in a correlator with a replica generated via a precision timegenerator. In order to obtain a replica like the encoding of the signalspicked up by the antenna, the generator is clocked by a clock signal ofa time base, and receives a code control signal from a code source. Atthe correlator output, the intermediate signals undergo temporalintegration prior to demodulation and summation of the pulses in orderto retrieve information from the received ultra-wide band signals.

One drawback of this apparatus is that a correlation operation has to becarried out prior to demodulating and adding the pulses of theintermediate signals to retrieve information. Moreover, the shape of thepulses must be known, and only the direct path signals or one of themultiple path signals can be detected with this apparatus, which is adrawback.

U.S. Pat. No. 6,483,461 discloses an ultra-wide band signal receptionapparatus, which includes the same elements as the apparatus in USPatent Application No. 2003/0095609 so as to be used for positioningpurposes. Consequently, the same drawbacks are noted as with thereception apparatus of the preceding Patent Application.

US Patent Application No. 2003/0058963 discloses a method and a devicefor receiving ultra-wide band type pulse signals. The signals include aheading frame for seeking synchronisation in the reception device. Inorder to do this, the ultra-wide band signals are received by an antennaof the device in order first of all to be compared to a thresholdvoltage in a comparator. At the output of the comparator, intermediatesignals represent the sign of the received signal in relation to athreshold voltage. These intermediate signals are then sampled insampling means, and sliding correlation is performed on a final set ofsamples using a reference replica to remove noise. This set of samplesresults of an addition of several groups of sampled signal samples. Eachgroup of samples represents one of the pulses of the ultra-wide bandsignals, whose the temporal width of each group is equal to or greaterthan the reverse of a pulse repetition frequency of the ultra-wide bandsignals.

One drawback of such a device is that the information relative to thesign of pulses of ultra-wide band signals has to be exclusively used.Furthermore, the synchronisation check has to be carried out by usinginformation after correlation operation. A pulse energy maximization inthe set of samples from the addition directly is not carried out, whichis another drawback.

US Patent Application No. 2003/0198308 discloses a UWB time referencedelay-hopped TR/DH communication system. The system reception deviceincludes several pulse pair correlators operating in parallel to performauto-correlation of the signals received by an antenna, and ananalogue-digital converter at the output of each correlator. Theinformation is subsequently demodulated using a known CDMA technique.

One drawback of this device is that it is necessary to carry outcorrelation operations as soon as the UWB signals are received, whichcomplicates the manufacture of this device in the same way as for USPatent Application No 2003/0095609. Further, the communication system islimited to double pulse signals.

US Patent Application No. 2003/0002347 discloses an ultra-wide bandsignal reception apparatus, which includes the same elements as theapparatus of US Patent Application No. 2003/0198308 so as to be used forpositioning purposes. Consequently, the same drawbacks are observed aswith the reception apparatus of the preceding Patent Application.

It is thus a main object of the invention to overcome the drawbacks ofthe prior art by providing a wireless data communication method viaultra-wide band encoded data signals which is able to process simply allof the encoded direct path and/or multiple path signals picked up by thereceiver device.

It is another object of the invention to provide a wireless datacommunication method via encoded ultra-wide band data signals formaximising the amplitude of the data pulses in relation to the noisepicked up by the receiver device.

The invention therefore concerns an aforecited method which ischaracterized in that the N pulses of a pulse sequence of encoded directpath and/or multiple path data signals received by the receiver deviceare each processed in one of N corresponding temporal reception windows,each of the N temporal reception windows being positioned in time as afunction of a known theoretical arrangement of the N pulses of thesignals transmitted by the transmitter device, and in that an operationof adding the N windows is performed in the receiver device such thatthe added pulse amplitude level is higher than noise amplitude levelpicked up by the receiver device.

One advantage of the communication method according to the invention isthat most of the pulses of the direct path and/or multiple path signalsreceived from each temporal window can be added coherently, since eachof the N temporal reception windows is positioned in time in accordancewith a known placing of the N encoded data signal pulses transmitted bythe transmitter device. Even if the direct path signals cannot be pickedup by the receiver device because of an obstacle on the signal path, itis possible to add coherently the pulses of each corresponding windowfrom the multiple path encoded signals.

This coherent addition of N windows does not occur in conventionalcommunication systems such as those disclosed in US Patent ApplicationsNos. 2003/0095609 and 2003/0198308.

Each window can be chosen with sufficient width to pick up each of the Npulses of all of the signals picked up by the receiver device. Thiswidth, which is the same for all the temporal windows, can be adjustedas a function of the propagation features of the transmission channel,and during the time and frequency search phase for received signal dataacquisition. The width of each window may be for example 20 or 50 ns,and each window is, as a rule, centred on a theoretic reference positionrelative to the direct path signal pulses.

The position of the start of each window corresponds to the position ofthe encoded data signal sequence pulses for carrying out coherentaddition of the pulses of each window. With this coherent addition ofthe temporal window pulses, the added pulse amplitude level becomeshigher than the noise level if the receiver device is properlysynchronised in time and frequency in relation to the transmitterdevice. After this temporal window addition step, data demodulation canbe carried out in a signal processing unit of the receiver device.

The window addition can be achieved analogically prior toanalogue-digital conversion of the data signals, or digitally afteranalogue-digital conversion. In order to reduce the electric powerconsumption of the receiver device, signal sampling can only be carriedout in an analogue-digital conversion stage during time intervals thatare identical to the duration of the temporal windows.

Because of the temporal window addition, the noise signal amplitudelevel picked up by the receiver device is greatly reduced, in relationto the added pulse level. This is due to the fact that the voltagepolarity of the noise signals in the time interval of each window is notprecisely defined, unlike the voltage polarity of the data signalpulses.

Preferably, the data is encoded by time modulation of the pulses of eachsequence as indicated hereinbefore. However, one could also envisageencoding the data by pulse polarity or phase modulation or by acombination of pulse time and polarity or phase modulation. In the caseof pulse polarity modulation, each window, which includes one or severalpulses, is multiplied by −1 for pulses of negative polarity and by +1for pulses of positive polarity so as to be able subsequently to add thepulses of all the windows in a coherent manner.

Another advantage of the communication method according to the inventionis that the clock or sampling signals of the receiver device can befrequency adjusted owing to the result of the temporal window addition.The clock or sampling signal frequency is adjusted to the clock signalfrequency of the transmitter device by a signal processing unit of thereceiver device. This frequency adjustment can be made at any time when,for example, an alteration to the position of the added pulses in afinal temporal window is observed, or when the added pulse amplitudelevel decreases.

In order to carry out this adjustment, the data signals transmitted bythe transmitter device can include a synchronisation frame. Thissynchronisation frame includes several successive sequences of Npersonalised pulses to the transmitter device. Thus, since the receiverdevice knows the position of the pulses of each of the sequences, it cancarry out a two dimensional time and frequency search to find the startof transmission and the frequency gap.

Owing to the communication method according to the invention, it ispossible to choose and track the sampling or clock signal frequency inorder to maximise the added pulse amplitude peak whether the pulses aredirect or multiple path signal pulses.

Another advantage of the communication method according to the inventionis that it can be used for positioning purposes. In order to do this, atleast two transmitter devices, or even three transmitter devices aregenerally required to transmit encoded data signals. This enables thereceiver device to calculate the positioning coordinates as a functionof the first encoded signal time of arrival as described hereinafter.For a positioning operation, the number N of pulses per transmitted datasequence can be for example equal to 1024 with a pulse width of forexample 0.5 ns.

Another advantage of the communication method according to the inventionis that the noise level picked up can be estimated in the signalprocessing unit of the receiver device. In order to do this, severalmaximum signal absolute value amplitudes are calculated successively orin parallel in one or several temporal sub-windows in the signalprocessing unit of the receiver device. These sub-windows are shifted byspecified time intervals from the start of the temporal window at theend of said temporal window. An estimation of the noise amplitude levelis carried out by selecting the minimum amplitude value from all thecalculated amplitude values. This estimation can be carried out beforeor after the temporal window addition operation.

Another advantage of the communication method according to the inventionis that it enables the time of arrival of the first direct path and/ormultiple encoded data signals to be calculated. In the case where thedirect path signals are not picked up by the receiver device, the firstmultiple path signals are processed. This time of arrival estimationoperation consists first of all in calculating a positive signalenvelope for each temporal window or the final temporal window.Afterwards, minimum and maximum points of the envelope are determinedand a central point is calculated, one of whose functions, which may betangent or linear, allows estimating the rising edge of the envelope.

The invention also concerns a receiver device for implementing thewireless data communication method wherein all of the direct path and/ormultiple path encoded signal pulses picked up can be processed simply.

Therefore, the receiver device for implementing the communicationmethod, which includes a second oscillator stage delivering at least asecond clock signal at a second defined frequency, a second signalprocessing unit connected to the second oscillator stage, and ananalogue-digital conversion stage for the encoded data signals receivedby the second wide band antenna, is characterized in that the signalprocessing unit includes temporal window adding means for coherentlyadding the pulses of each of the N temporal windows.

The objects, advantages and features of the wireless data communicationmethod via ultra-wide band signals, and of the receiver device for theimplementation thereof will appear more clearly in the followingdescription of embodiments of the invention with reference to theannexed drawings, in which:

FIG. 1 a shows schematically a data communication system forimplementing the communication method according to the invention,wherein the temporal windows are added digitally in a receiver device,

FIG. 1 b shows schematically a data communication system forimplementing the communication method according to the invention,wherein the temporal windows are added analogically in a receiverdevice,

FIG. 2 shows schematically how the signals of the N temporal windows areadded in a receiver device for the communication method according to theinvention,

FIGS. 3 a to 3 d show graphs showing a temporal data encodingmodulation, polarity data encoding modulation, temporal and polaritydata encoding modulation, and amplitude encoding modulation of thetransmitted data of the communication method according to the invention,

FIG. 4 shows in a simplified manner the encoded data signals startingwith a synchronisation frame for the communication method according tothe invention,

FIG. 5 shows signal graphs in the transmitter device and in the receiverdevice in the case of time and frequency synchronisation of the clocksignals of the two devices, in the case of a frequency gap between theclock signals and in the case of non temporal synchronisation of thewindows of the communication method according to the invention,

FIGS. 6 a and 6 b show one embodiment of an analogue-digital conversionstage of the receiver device, and clocking signals of the conversionstage for implementing the communication method according to theinvention,

FIG. 7 shows a signal graph of one temporal window of the receiverdevice of the steps for estimating the noise level picked up of thecommunication method according to the invention,

FIG. 8 shows a graph of one part of a temporal window of the receiverdevice of the steps for calculating the positive signal envelope of thetemporal window of the communication method according to the invention,and

FIG. 9 shows a graph of one part of a temporal window of the receiverdevice of the steps for calculating the time of arrival of the firstencoded direct path or multiple path data signals of the communicationmethod according to the invention.

In the following description, those elements of the wireless datacommunication system via ultra-wide band encoded data signals used forimplementing the communication method, which are well known to thoseskilled in the art, will not be explained in detail.

FIGS. 1 a and 1 b shows schematically a communication system 1 forimplementing the wireless data communication method via ultra-wide bandencoded data signals S_(D). Communication system 1 includes at least onetransmitter device 2 which transmits encoded data signals S_(D) via afirst wide band antenna 27 and a receiver device 3 which receives directand/or multiple path encoded data signals via a second wide band antenna37.

As explained hereinafter, particularly with reference to FIG. 2, thedirect path and/or multiple pulses received by receiver device 3 andcorresponding to transmission of one of the N encoded data signal pulsesare processed or selected in one of the N corresponding temporal windowsin receiver device 3. Since each temporal window is positionedchronologically as a function of the known position of each pulse of thetransmitted encoded data signals S_(D), an addition of the temporalwindows is performed in order to add coherently the pulses of eachwindow.

Generally, transmitter device 2 includes a stage oscillator 21 forproviding a clock signal CLK_(e), whose frequency depends upon a quartzresonator 22, a signal processing unit 23 clocked by the clock signal,and a pulse shaping unit 24 connected to signal processing unit 23.Given the use of a stage oscillator 21 with a quartz 22, the frequencyof clock signals CLK_(e) can preferably be multiplied M times in signalprocessing unit 23. This multiplication by M of the frequency of clocksignals CLK_(e) is obtained conventionally using lag gates that are notshown, and a combination of the clocking pulses at output of the laggates.

In signal processing unit 23, the useful frequency for generating datapulses may be greater than or equal to 1 GHz. This requires the use ofat least 4 lag gates shifted by a quarter period in relation to a periodof clock signal CLK_(e) at a frequency of the order of 250 MHz.

For UWB encoded data signal transmission, processing unit 23 oftransmitter device 2 has to provide, to pulse shaping unit 24, one orseveral sequences of N successive pulses of positive or negative voltageor current polarity. Each pulse of the sequences is generated in a timeinterval corresponding to the reverse of a pulse repetition frequency.For UWB data signals, this pulse repetition frequency (PRF) can behigher than or equal to 10 MHz.

The way in which the data is encoded in sequences of N pulses in signalprocessing unit 23 of transmitter device 2 must, on the one hand,differentiate each symbol or character to be transmitted and, on theother hand, personalise the transmitter device. A close receiver device3 can thus recognise where the received data signals have come from,since the various codes used for personalising the transmitter devicesare orthogonal.

Pulse shaping unit 24 receives the data in the form of one or moresequences of N pulses to be transmitted by the first UWB antenna 27 ofsignal processing unit 23. These encoded data pulses in processing unit23 are amplified in an amplifier 25 of pulse shaping unit 24, andfiltered in a conventional bandpass filter 26 prior to being transmittedby first UWB antenna 27. Generally, the shape of the energy pulses ofdata signals S_(D) transmitted by first UWB antenna 27 is obtained byderivation of the shaped pulses, due to an antenna current switch. Thetransmitted pulses must be Gaussian shaped with one or two alternations,or of another shape.

FIGS. 3 a to 3 d show the way in which the data is encoded, such as oneor more characters or symbols, by one or more sequences of N pulses inthe data signals.

The data can be encoded by temporal modulation of the pulses of eachsequence, as shown in FIG. 3 a. This is called Pulse Position Modulation(PPM). The pulses presented are Gaussian shaped with two alternations.Of course, the pulse shape can also be Gaussian with one positive ornegative alternation, or of various other shapes.

In FIG. 3 a, each character C1 and C2 is defined by N pulses, each pulseof which is of smaller length than 1 ns, in a sequence repetition periodT_(rép). Each pulse is generated by temporal interval 1/PRFcorresponding to the reverse of the pulse repetition frequency PRF asdescribed hereinbefore. The temporal position of each pulse in thetemporal interval is specific to the character to be encoded. Moreover,the gap between each pulse of the sequence of N pulses is preferablypseudo-random to personalise the transmitter device. With thisarrangement of N pulses per sequence repetition period T_(rép), thecharacter or symbol C2 differs from character or symbol C1 only by atemporal difference dt of each of the N pulses generated. Of course, forother characters or symbols to be transmitted, the temporal differencedt is different each time.

The sequence repetition time T_(rép) can be for example 0.1 ms with 1024pulses per sequence, or 10 μs with 256 pulses per sequence.

The data can also be encoded by polarity or phase modulation of thepulses generated by the signal processing unit of the transmitter deviceas shown in FIG. 3 b.

In this FIG. 3 b, it will be noted that the identical temporaldifference between each pulse is equal to a period repetition value1/PRF. Conversely, the pulse polarity, particularly, the phase, is anencoding feature that personalises the transmitter device, as is thecharacter or symbol D1 or D2 to be transmitted in the data signals.

The positive polarity or zero phase of double alternation pulses candefine a +1 state, whereas negative polarity or 180° phase of doublealternation pulses can define a −1 state. Since the shape of the pulsesshown in FIG. 3 b is Gaussian with two alternations, the differencebetween a +1 state and a −1 state is observed via a 180° phase shift ofthe pulse. However, one could very well have imagined a Gaussian shapewith positive alternation to define a +1 pulse state, or a negativealternation to define a −1 pulse state.

In FIG. 3 c, the data is encoded by a combination of temporal andpolarity modulation of the pulses. The N pulses of a sequence fordefining the character or symbol E1 or E2 are presented with a simplealternation. Each pulse can be of positive or negative polarity.However, the character or symbol E2 differs from character or symbol E1via a temporal gap dt of each generated pulse. It should be noted thatthe polarity of each 1 alternation pulse of each character could also bedifferent.

Finally, FIG. 3 d shows data encoding via amplitude modulation of thesimple positive alternation pulses. The amplitude of a pulse below adetermined amplitude threshold defines a 0, whereas the amplitude of apulse above a determined threshold defines a 1. In the case of pulseamplitude modulation, the identical temporal gap between each pulse isequal to a pulse repetition frequency value 1/PRF. The character orsymbol F1 differs from character or symbol F2 by a sequence of N pulsesof different amplitude.

It should be noted that pulse amplitude modulation is not a robustmethod. Moreover, it is difficult to implement in UWB technology, whichmeans that preferably, the data is encoded in accordance with one of themodulation methods shown in FIGS. 3 a to 3 c, or a combination of thesemodulation methods.

For the reception of direct path and/or multiple path encoded datasignals S_(D), receiver device 3 includes first of all a second wideband antenna 37. This antenna 37 provides signals, which are derived onthe basis of the picked up encoded data signals, to a low noiseamplifier (LNA) 36, fitted with a band pass filter. After this LNA 36,an automatic gain control amplifier (AGC) 35 can be provided, whoseamplification factor A_(C) is controlled by control means 43 of a signalprocessing unit 33. Amplifier 35 provides amplified intermediate signalsS_(INT) to an analogue-digital conversion stage 34 responsible for thedigital conversion of the analogue signals.

Receiver device 3 further includes a stage oscillator 31 for supplying aclock signal CLK_(r), whose frequency depends upon a quartz resonator32, and a signal processing unit 33 clocked by clock signal CLK_(r).Clock signals CLK_(r) are provided in particular to the signalprocessing unit control means 33.

Given the use of a stage oscillator 31 with a quartz 32, control means43 are responsible for multiplying clock frequency CLK_(r) by a factor nas for the transmitter device described hereinbefore. On the basis ofthe clock signals CLKr, control means 43 provide in particular clockingsignals CLK_(1-n) to analogue-digital conversion stage 34 for samplingoperations. This conversion stage 34 will be described hereinafter withreference to FIGS. 6 a and 6 b.

It should be noted that in order to reduce the electrical powerconsumption of the receiver device, one could envisage only sampling theintermediate signals during periods identical to the temporal width ofeach window.

According to a first embodiment of receiver device 3 of FIG. 1 a, signalprocessing unit 33 includes in addition to control means 43, digitalwindow addition means 41 for receiving sampled signals S_(NUM) from theanalogue-digital conversion stage 34, data demodulation means 42 andtime of arrival estimation means 44. Means 42 and 44 are both connectedto the output of digital window addition means 41 for receiving signalsfrom a final addition window W_(S).

In order to control the operations of signal processing means 33,control means 43 first of all provide control signal C_(FN) to digitalwindow addition means 41. These control signals C_(FN) adjust thetemporisation of the temporal selection windows of parts of the digitalsignals, i.e. the placing of the first of the N windows in time.

In order to arrange the temporal windows, a two dimensional time andfrequency search must therefore be carried out. This search will provideproper synchronisation and a clock frequency of oscillator stage 31proportionally adapted to the clock frequency of oscillator stage 21,which is the basis of the generation of the transmitted encoded datasignal pulses. Thus, control means 43 can directly adjust the frequencyof clock signals CLK_(r) by control signals C_(H). These control signalsC_(H) can adapt a resistive or capacitive value of a network of wellknown resistors or capacitors of oscillator stage 31.

Another frequency search method consists in using control signals C_(FN)to alter the pulse time or repetition frequency scale of the N windowsto be added of the digital window addition means 41. This meansperforming a re-sampling operation in signal processing unit 33 ofreceiver device 3 with a different re-sampling frequency from thesampling frequency of analogue-digital conversion stage AN 34. There-sampling frequency generated by control means 43 may be much higherso as to increase precision particularly for positioning.

Once the window addition operation has been performed in digital windowaddition means 41, the control means supply control signals C_(D) to thedata demodulation means 42. These data demodulation means are able toprovide data only if the N pulses of a temporal window sequence havebeen coherently added.

In order to recognise the character(s) or symbol(s) transmitted in theencoded data signals, signals W_(S) of the final window must present oneor several pulses to demodulation means 42 whose amplitude is higherthan a determined threshold and at the noise level picked up by receiverdevice 3. In this way, it is possible to determine the character(s) orsymbol(s) particularly by the position of the pulses in the final windowfor PPM type modulation.

It should be noted that the maximum amplitude pulse of the final windowis not necessarily due to the N added pulses of the direct path signals,since it is possible for obstacles on the path of the encoded datasignals, to attenuate the amplitude of each direct path signal pulse orto prevent reception of such signals. However, since the N pulses of allthe direct or multiple path encoded data signals have each to beprocessed in one of the N width-adapted temporal windows, it is possibleto provide final window signals W_(S) to demodulation means 42 in whichat lest one maximum pulse results from multiple path signals.

In order to estimate the noise level and time of arrival of the pulsesof the first direct or multiple path encoded data signals, control means43 provide control signals C_(E) to time of arrival estimation means 44so that time of arrival data TOA is provided. These time of arrivalestimation means are explained hereinafter with reference to FIGS. 7 to9.

According to a second embodiment of the receiver device presented inFIG. 1 b, the essential difference in relation to the first embodimentof FIG. 1 a is that the window addition occurs in analogue windowaddition means 45. These means 45 can be outside signal processing means33 or incorporated therein. The analogue window addition means 45 can beinserted between amplifier 36 and amplifier 35. However, means 45 canalso be placed after amplifier 35 and before analogue-digital conversionstage 34.

Conventionally, in order to add up all the temporal windowsanalogically, a number N-1 of temporisation gates are used, not shown,whose time period is adjusted to the position of each of the desired Nwindows. The encoded data signals received by antenna 37 pass througheach of these gates so as to be able to add up in proper synchronism,for a time period equivalent to the width of each window, the outputsignals of each temporisation gate and the input signals of the first ofsaid gates. The signals resulting from this addition are then amplifiedby amplifier 35 and sampled by conversion stage 34.

Analogue-digital conversion stage 34 provides digital signals S_(NUM)matching the sampling of signals from the final addition window ofanalogue window addition means 45. These digital signals S_(NUM) aredirectly processed by demodulation means 42 and time of arrivalestimation means 44.

FIG. 2 shows the temporal window addition operation which is a mainfeature of the data communication method whether the addition is ofanalogue or digital signals.

The encoded data signals, which are picked up by the second antenna ofthe receiver device, include noise in addition to the pulses of eachsequence defining the data to be demodulated. This noise is representedin FIG. 2 by dotted lines to distinguish it from the encoded data signalpulses. It can be seen that in each window FEN₁ to FEN_(N), direct andmultiple path signal pulses are picked up by the receiver device, butwith a lower amplitude level than the noise level.

The N windows, which contain the pulses of all the picked up encodeddata signals, originating from a specific transmitter device, arearranged in accordance with a time arrangement determined as a functionof the known theoretical position of each direct path encoded datasignal pulse. The width of each window T_(W) is adapted so as to be ableto detect the pulses of several direct and multiple path encoded datasignals bearing the same data, which is one advantage of the presentinvention.

Each temporal window can have a width comprised between 20 and 50 ns forexample, and starts before the appearance of each pulse of the directpath signals. However, this width may be smaller while picking up atlast one of the multiple path signals in addition to the direct pathsignals, or also larger for example of the order of 100 ns in the caseof positioning.

In a positioning or text or synchronisation data communication system,it is generally advantageous for the width of the temporal windows to belarger during the temporal synchronisation search. This enables directand/or multiple path signals to be detected which may be partiallyreceived with a lot of lag or lead on the theoretic desired position.

When temporal synchronisation is found and the clock signal frequency ofthe transmitter and receiver devices has been properly adjusted, eachpulse of a data sequence is properly located in each temporal window.Consequently, when all of the temporal windows FEN1 to FEN are added upby at least one adder 51, all of the pulses of sequences of all thesignals picked up by the receiver device are added up coherently tomaximize the pulse amplitude level in relation to the noise level. Sincethe noise signal voltage polarity is not precisely defined in the timeinterval of each window, unlike the voltage polarity of the data signalpulses, after the addition operation the noise amplitude level is lowerthan the pulse amplitude level.

In order to obtain coherent addition of the pulses of each window, theremust be proper synchronisation between the transmitter device and thereceiver device. In order to do this before transmitting variouscharacters or symbols in the data signals, one may wish to transmit asynchronisation frame at the start as shown symbolically in FIG. 4. Thissynchronisation frame is composed of one or several successive sequencesof N pulses. This leaves time for the receiver device to adjust theplacing of the windows as a function of the position of each pulse ofthe sequences. Moreover, this leaves time for synchronising thefrequency of the second oscillator stage 31 with first oscillator stage21, or synchronising the re-sampling frequency of digital windowaddition means 41.

In order to understand the importance of having proper synchronisationbetween the transmitter device and the receiver device so as to be ableto demodulate the received encoded signal data, reference can be made tovarious signals shown in FIG. 5. Signals A to C are transmitter devicesignals, whereas signals D to I are receiver device signals. In thisFIG. 5, the number N of pulses is chosen to be equal to 5 which matchesa processing gain PG of the order of 7 dB after the temporal windowaddition operation in the receiver device.

This processing gain can be calculated using the formula PG=10·logN[dB], which means that if a larger gain is desired, each sequence,which defines one or several characters, must include a large number Nof pulses. Of course, with a larger number of pulses per sequence, it isinevitable that data demodulation will slow down, but this may betolerated depending upon the type of data to be transmitted. Forexample, with a number N equal to 200, the processing gain will be ofthe order of 23 dB, and with a number N equal to 1024, the processinggain will be of the order of 30 dB.

Signals A are reference clock signals with a frequency f₀ which are usedfor clocking data modulation in the processing unit of the transmitterdevice.

Signals B are signals leaving the processing unit of the transmitterdevice which include one rectangular pulse per pulse repetition period1/PRF. These signals B are trigger signals for the pulse shaping unit ofthe transmitter device.

Signals C are encoded data signals transmitted by the wide band antennaof the transmitter device. The data in these encoded signals are definedby double alternation pulses of smaller width than 1 ns.

Signals D are signals picked up by the wide band antenna of the receiverdevice. It will be noted that these signals can contain direct pathand/or multiple path pulses, which can have a different shape from thepulses transmitted after the wide band antenna of the receiver device.In practice, a drift can be observed in the encoded signal pulses.

Signals E are clocking signals for sampling the analogue signals in theanalogue-digital conversion stage of the receiver device. The samplingfrequency f_(s) of signals E is identical to the frequency f_(s) of thereference clock signals of the transmitter device.

Signals F are clocking signals for sampling the analogue signals in theanalogue-digital conversion stage of the receiver device, whose samplingfrequency f_(s) has a frequency drift df relative to frequency f. Thisfrequency has to be adjusted in the receiver device during the twodimensional time and frequency search phase.

Signals G are temporal windows of samples of selected parts of the datasignals where the time between each start of a window exactly matchesthe time between each pulse of the data sequence. The sampling frequencyf_(s) is adjusted to the frequency f₀ of the reference clock signals asshown by signals E. When the pulses of each of these windows of widthT_(W) are coherently added up in the receiver device, it will be notedthat the pulse amplitude level becomes higher than the noise level infinal window G_(F).

It should be noted that each temporal window receiving a part of theencoded data signals can be obtained, in the data processing unit of thereceiver device, by a multiplication by 1 of the encoded signal parts tobe selected, and by 0 of the parts to be removed.

Signals H are temporal windows of samples of selected data signal partswhere a clock frequency drift is observed between the transmitter deviceand the receiver device by using sampling signals, like signals F. Inthis case, addition of the pulses of each window does not provide anadded pulse amplitude level that is higher than the noise level in finalwindow H_(F).

Signals I are temporal windows of samples of selected data signal partswhere the time between each start of a window exactly matches the timebetween each pulse of the data sequence, but without temporalsynchronisation between the transmitter device and the receiver device.The sampling frequency f_(s) is however well adjusted to the referenceclock frequency f₀ as shown by signals E. However, the start of the Nwindows is shifted time-wise, which means that no sequence pulse ispicked up by the receiver device and gives a window addition without anypulses as shown in the final window I_(F).

Since the receiver device knows the arrangement of the pulse sequencesto be picked up, a first step consists in finding the start of eachpulse sequence, either by time shifting in a serial manner or bysearching in parallel at several different times. If the samplingfrequency is not sufficiently close to the reference clock signalfrequency of the transmitter device, this search can be repeated eitherin series, or in parallel with different sampling frequencies.

Once temporal synchronisation has been found, the sampling orre-sampling frequency can be adapted to the reference clock signalfrequency of the transmitter device by controlling the pulse amplitudelevel in the final window until this amplitude level is maximisedrelative to the noise level.

At any time, the sampling or re-sampling frequency can be adapted bycontrolling any decrease in the pulse amplitude level in the finaltemporal window or by progressively moving pulses in said final window.The movement of added pulses in the final temporal window can be due tothe Doppler effect if the transmitter device moves away from or towardsthe receiver device.

If the temporal window addition is performed analogically, as describedhereinbefore and illustrated in FIG. 1 b, the sampling frequency will befrequency CLK_(r), which controls the analogue window addition.

FIGS. 6 a and 6 b show an embodiment of the analogue-digital conversionstage of the receiver device, and the clocking signals of the stageconverters.

The analogue-digital conversion stage includes a number n of convertersAN 53 to 55 working in parallel. Each converter 53 to 55 is clocked by aclocking signals CLK₁, CLK₂ to CLK_(n) with an identical frequency tothat of clock signals CLK_(r) generated by the oscillator stage. Eachclocking signal CLK₁, CLK₂ to CLK_(n) is phase shifted by 360°/n foreach converter 53 to 55. Consequently, the n phase shifted clockingsignals allow sampling of intermediate analogue signals S_(INT) at aneffective frequency f_(e) of n times the frequency of clock signalsCLK_(r).

Since the intermediate signal sampling is generally carried out at afrequency of 2 times the band width of the encoded data signals, forexample at a frequency that may be equal to or higher than 2 GHz, onecould envisage having 4 converters clocked by 4 clocking signals phaseshifted in relation to each other by 90° as illustrated in Figure B. Thefrequency of clock signals CLK_(r) must thus be 4 times less than theeffective sampling frequency f_(e).

At each rising edge of the clocking signal, each converter 53 to 55provides binary m bit signals S_(D1) to S_(Dn), where m can have a valuefrom 1 to 8. These binary signals S_(D1) to S_(Dn) are provided toseries input and parallel output type combination means 56, which areresponsible for combining all of the signals received from theconverters in order to provide digital signals S_(NUM) for the signalprocessing unit of the receiver device.

FIGS. 7 to 9 show graphs of signals of a temporal window for estimatingnoise level, calculating the signal envelope of a window and the time ofarrival of the data signals. These operations are carried out in thetime of arrival estimation means 44 of signal processing means 33, shownin FIGS. 1 a and 1 b, under the control of control signals C_(E)generated by control means 43.

First of all, FIG. 7 shows a method of estimating noise level A_(N)using a graph of the signals of one temporal window. This method isbased on the fact that inside the temporal window observation interval,there is at least one temporal window portion of length T_(N) duringwhich there is not energy belonging to the transmitted data sequencepulses. The estimated noise level A_(N) is lower than the maximumamplitude level A_(P) of the coherently added pulses.

In order to estimate noise level A_(N), several absolute value maximumamplitude values A_(i) are calculated for signals s_(i)(t), with iranging from 0 to I, in temporal sub-windows of length T_(N). The I+1temporal sub-windows for calculating the amplitude values are timeshifted in relation to each other by a determined time interval from thestart of the temporal observation window to the end of said temporalwindow. For I+1 temporal windows to be calculated, the number of timeintervals is I.

The absolute value noise amplitude level value A_(N) is equal to theminimum amplitude value among the A_(i) calculated values, or to theminimum value of the maximum of all the signals s_(i)(t).

FIG. 8 shows a method of calculating the positive envelope of thedigitised temporal window signals over a part of said temporal window.

According to this method, firstly, all of the zero crossing positionsp_(i) of the temporal window signals are determined, i.e. all thepositions where sampling before and after p_(i) has an opposite sign.After this step, the coordinates (x_(i), y_(i)) of the absolute valueamplitude maximum is determined in each interval from p_(i) to p_(i+1),with i ranging from 1 to I-1. Afterwards, the envelope is calculatedusing an interpolation algorithm which may be for example the piecewisecubic Hermite interpolation algorithm.

Finally, with reference to FIG. 9, there is shown a method of estimatingthe time of arrival of the first data signals received by the receiverdevice. These first signals may be direct path signals or multiple pathsignals in the absence of any direct path.

For this estimation, an amplitude threshold th is first calculated whichis based on the envelope amplitude peak A_(P), and on the noiseamplitude level estimation A_(N) described with reference to FIG. 7.This threshold th may be calculated by the following formula:th=5·A_(N)+A_(P)/25.

Next, the rising edge of the envelope where threshold th is exceeded forthe first time is estimated, by selecting a segment of the envelopeshown in bold in FIG. 9. An approximation of this segment with a givenfunction is carried out so that it can be used to estimate the risingedge of the envelope. In order to do this, the maximum local point ofthe envelope is first estimated at the coordinates (x_(M), y_(M)) whichdirectly follow the point where the envelope passes above threshold th.The minimum local point of the envelope is also estimated at thecoordinates (x_(m), y_(m)) that precede the point where the envelopepasses above threshold th.

After establishing these coordinates, the value y_(h), which is equal to0.5·(y_(M)+y_(m)) is calculated, which allows the correspondingcoordinate x_(h) to be found. A time value t₁≦min(x_(M)−x_(h),x_(h)−x_(m)) can then be selected.

After having selected time value t₁, a selection is made of a samplesequence from the envelope of length 2·t₁ centred on coordinate x_(h).Finally, an approximation is made of the selected sample segment of theenvelope with a given function in a least square direction. Thisfunction may be of the linear type, which allows the rising edge of theenvelope to be estimated based on this function. At point y=0 of thislinear function, the time of arrival of the first signals can thus bedetermined.

From the description that has just been given, those skilled in the artcan devise multiple variants of the data communication method via pulsesignals without departing from the scope of the invention defined by theclaims. The receiver device may not have an integrated filter low noiseamplifier, since the wide band antenna of the receiver device canalready fulfil the filtering functions. The receiver device can bearranged to act as the transmitter device, and the transmitter devicemay be arranged to act as the receiver device so that a data exchangecan occur.

1-17. (canceled)
 18. Wireless data communication method between atransmitter device having a first wide band antenna for transmittingultra wide band coded data signals, and a receiver device having asecond wide band antenna for receiving direct path and/or multiple pathcoded data signals, the transmitted data being defined by one or moresequences of N pulses where N is an integer number higher than 1, thearrangement of N pulses of each sequence representing encoding of datarelating to the transmitter device, wherein the N pulses of one pulsesequence of direct path and/or multiple path coded data signals receivedby the receiver device are each processed in one of N correspondingreception time windows, each of the N reception time windows beingpositioned in time as a function of a known theoretical arrangement ofthe N pulses of the signals transmitted by the transmitter device, andwherein an operation of adding the N windows is carried out in thereceiver device so that the added pulse amplitude level is higher thanthe noise amplitude level captured by the receiver device. 19.Communication method according to claim 18, wherein a clock signalfrequency for clocking various operations of the receiver device isproportionally adapted to a reference clock signal frequency of thetransmitter device, which is used for generating ultra-wide band codeddata signals, by controlling the pulse amplitude level of a final windowadding the N windows until said amplitude level is maximised. 20.Communication method according to claim 18, wherein the transmitterdevice transmits coded data signals, in which the data is coded by pulseposition modulation of each sequence, or by pulse polarity or phasemodulation of each sequence, or by pulse position and polaritymodulation of each sequence.
 21. Communication method according to claim18, wherein the coded data signals include a synchronisation frameallowing the receiver device to recognise the transmitter device and tobe synchronised on said frame before demodulating the received data,said synchronisation frame being composed of one or several sequences ofN pulses of determined pulse repetition frequency.
 22. Communicationmethod according to claim 18, wherein the identical width of each of theN time windows is smaller than the reverse of the mean pulse repetitionfrequency of a sequence of coded data signals to be transmitted, andwherein said time window width is adapted to receive the pulses of thedirect path and multiple path signals captured by the receiver device,for example of width greater than 20 ns.
 23. Communication methodaccording to claim 18, wherein the transmitter device includes a firstoscillator stage delivering at least one first clock signal at a firstdefined frequency, a first signal processing unit clocked by the clocksignal provided by the first oscillator stage in order to modulate thedata to be transmitted, and a unit for shaping the N pulses of eachsequence to be transmitted by the first wide band antenna of thetransmitter device as a function of the data modulation provided by thefirst signal processing unit, wherein the receiver device includes asecond oscillator stage delivering at least one second clock signal at asecond defined frequency, a second signal processing unit connected tothe second oscillator stage, and an analogue-digital conversion stagefor analogue signals relating to the coded data signals received by thesecond wide band antenna, wherein an operation of adding the N timewindows is performed before or after the analogue-digital conversion ofthe analogue signals, and wherein the analogue signal pulses are sampledin the analogue-digital conversion stage by at least one sampling signalsupplied by the second signal processing unit, the sampling signalhaving a frequency proportional to the second frequency of the secondclock signal.
 24. Communication method according to claim 23, whereinthe time window signals are successively added and stored in at leastone register of the second signal processing unit.
 25. Communicationmethod according to claim 19, wherein each reception window positionedin time in relation to the known theoretical place of each pulse of thereceived data signals is centred relative to a theoretical referencevalue or relative to the maximum added pulse amplitude of the directpath and/or multiple path signals captured by the receiver device. 26.Communication method according to claim 20, wherein the referencesignals of identical polarity to the polarity of the coded signalsreceived by the receiver device are correlated prior to addition of theresulting pulses of each time window.
 27. Communication method accordingto claim 23, wherein the second signal processing unit includes meansfor adding the digital windows and means for estimating the time ofarrival of the coded data signals, wherein before or after the timewindow addition operation, the method includes steps consisting incalculating several absolute value maximum amplitude values for signalsin time sub-windows of defined length T_(N), each of the sub-windowsbeing time shifted in relation to each other by a determined timeinterval from the start of the reception time window to the end of saidtime window, and in estimating a noise amplitude level by selecting theminimum amplitude value from all the calculated amplitude values. 28.Communication method according to claim 23, wherein it includes stepsfor calculating a positive envelope of the signals of one time windowconsisting in determining all the zero crossing positions p_(i) of thetime window signals, in determining the coordinates of the absolutevalue amplitude maximum in each interval from p_(i) to p_(i+1), with iranging from 1 to I-1, I being an integer number higher than 3, and incalculating the envelope by using a specific interpolation algorithmpassing through the determined coordinates.
 29. Communication methodaccording to claim 28, wherein it includes steps for calculating thetime of arrival of the first signals captured by the receiver deviceconsisting in calculating an amplitude threshold th based on theamplitude maximum of the envelope, and an estimated noise amplitudelevel, in estimating the rising edge of the positive envelope where thethreshold th is exceeded for the first time, in estimating the maximumlocal point of the envelope at the coordinates which directly follow thepoint where the envelope passes above the threshold th, and the minimumlocal point of the envelope at the coordinates which precede the pointwhere the envelope passes above the threshold th, in calculating theintermediate coordinates between the minimum point and the maximumpoint, in approximating at the position of intermediate coordinates aselected segment of samples of the envelope with given function, such asan affine function, and in determining the time of arrival of the firstsignals captured by the receiver device at the zero crossing or anothervalue of the determined function.
 30. Communication method according toclaim 23, wherein the second signal processing unit includes controlmeans for providing control signals to digital window addition means inorder to modify the time or mean repetition frequency scale of N windowsto be added from digital window addition means, wherein a re-samplingoperation is carried out in the second signal processing unit of thereceiver device with a different re-sampling frequency from the samplingfrequency of the analogue-digital conversion stage, said re-samplingfrequency generated by the control means being able to be higher thanthe sampling frequency in order to increase precision for positioningpurposes.
 31. Receiver device for implementing the communication methodaccording to claim 18, including an oscillator stage delivering at leastone clock signal at a defined frequency, a signal processing unitconnected to the oscillator stage, and an analogue-digital conversionstage for the coded data signals received by a wide band antenna,wherein the signal processing unit includes time window addition meansfor coherently adding up the pulses of each of the N time windows. 32.Receiver device according to claim 31, wherein the clock signalfrequency of the oscillator stage is proportionally adapted by theprocessing unit to a reference clock signal frequency of an oscillatorstage of the transmitter device, which is used for generating ultra-wideband coded data signals, by controlling the pulse amplitude level of afinal addition window of the N windows from the addition means untilsaid amplitude level is maximised.
 33. Receiver device according toclaim 31, wherein the time window addition means receive digital signalsfrom the analogue-digital conversion stage for adding up the digitalwindows.
 34. Receiver device according to claim 31, wherein the timewindow addition means receive analogue data signals from the second wideband antenna in order to add up the analogue windows.